According to a switching power supply device in which an alternating voltage from an AC source is converted into a direct voltage by a rectifier and a first smoothing capacitor, a waveform of an electric current supplied to the switching power supply device is distorted. This causes a reduction in a power factor.
In view of this, a conventional switching power supply device includes a circuit which suppresses the distortion of the waveform of the electric current supplied to the switching power supply device. In such a circuit, a boost chopper circuit is connected with an output terminal of the rectifier. The boost chopper circuit is constituted by a boost inductor, a switching element, a rectifier diode, and a second smoothing capacitor. Such a circuit is called a power factor improvement circuit because the circuit improves a power factor by suppressing a distortion of a current waveform.
There have been two methods for controlling the power factor improvement circuit. A first method is a DCM (Discontinuous Conduction Mode). According to the DCM, the switching element is caused to be in an ON state for a certain period of time so that an electric current passes through the boost inductor. When the switching element goes into an OFF state, a state in which no electric current is passing through the boost inductor is detected, and the switching element is again caused to be in the ON state. A second method is a CCM (Continuous Conduction Mode). According to the CCM, a PWM (Pulse Width Modulation) control is carried out at predetermined intervals regardless of the electric current passing through the boost inductor.
FIG. 5, which corresponds to FIG. 1 of Patent Literature 1, is a circuit diagram illustrating a conventional switching power supply device 101 including a power factor improvement circuit. The power factor improvement circuit of FIG. 5 employs the CCM. The switching power supply device 101 of FIG. 5 includes (i) a filter 111 which filters out noise contained in an AC input voltage Vin and (ii) a bridge rectifier circuit DB which rectifies the AC input voltage Vin supplied via the filter 111. The switching power supply device 101 further includes a smoothing capacitor C101 which smoothes a rectified voltage supplied from the bridge rectifier circuit DB.
The smoothing capacitor C101 is connected, at its both terminals, with a first series circuit which includes a boost inductor L101, a switching element Q101, and a resistor R104. The switching element Q101 is for example a MOSFET (Metal Oxide Semiconductor Field Effect Transistor). Between a drain and a source of the switching element Q101, a second series circuit, which includes a rectifier diode D101 and a smoothing capacitor C104, is connected. The smoothing capacitor C104 is connected, at its both terminals, with a third series circuit which includes a resistor R108 and a resistor R109.
According to the conventional switching power supply device 101 which includes the power factor improvement circuit employing the CCM, in a case where electric power greater than or equal to a certain value is supplied to a load, a direct current is superimposed on an electric current passing through the boost inductor L101. Accordingly, an electric current constantly passes through the boost inductor L101.
Note here that the electric power greater than or equal to a certain value depends on (i) inductance of the boost inductor L101, (ii) a period during which the switching element Q101 is in the ON state, (iii) a voltage applied to the boost inductor L101, and the like.
When the direct current is superimposed on the electric current passing through the boost inductor L101, the switching element Q101 goes into the ON state while an electric current is flowing from the boost inductor L101 to the rectifier diode D101. This causes a state of the rectifier diode D101 to abruptly change from an ON state to a state in which a voltage in an opposite direction is applied across the rectifier diode D101. That is, after the switching element Q101 goes into the ON state, a recovery current passes through the rectifier diode D101 from a cathode to an anode.
The recovery current is a pulsed current which flows for a short period of time. However, the recovery current is larger than an electric current passing through the rectifier diode D101 while the rectifier diode D101 is being in the ON state. Therefore, the recovery current causes noise. In order to suppress noise, generally, a snubber circuit is provided so as to be connected in parallel with the rectifier diode D101.
The conventional power factor improvement circuit of FIG. 5 includes a fundamental wave component extraction circuit 118 provided between (i) a gate of the switching element Q101 and an output terminal of an AND circuit 127 and (ii) a resistor RT (timing resistor) of an oscillator 119.
The fundamental wave component extraction circuit 118 includes (i) a resistor R110 and a resistor R111 which are connected in series with each other and (ii) a capacitor C105. The capacitor C105 has (a) a first terminal which is connected to a connection point between the resistor R110 and the resistor R111 and (b) a second terminal which is electrically grounded.
The oscillator 119 is capable of changing an oscillation frequency by a control signal G supplied from the fundamental wave component extraction circuit 118, which control signal G is a fundamental wave component of a control signal F supplied to the gate of the switching element Q101. As the oscillation frequency of the oscillator 119 changes according to a change in the control signal G, a frequency (i.e., duty) of the control signal F (a PWM signal) changes.
As the frequency of the control signal F (the PWM signal) changes like above, frequencies of harmonic components of the control signal F are dispersed. As the frequencies of the harmonic components are dispersed, frequencies of noise caused by the recovery current are also dispersed. Accordingly, superimposition of noise at an identical frequency does not occur, and thus the noise can be reduced (that is, it is possible to further reduce a level of noise, refer to Patent Literature 1).
According to the switching power supply device 101 of FIG. 5, the bridge rectifier circuit DB rectifies the AC input voltage Vin via the filter 111 which serves as a noise filter. A rectified voltage outputted from the bridge rectifier circuit DB is then supplied to the boost chopper circuit via the smoothing capacitor C101 (normal mode filter).
The boost chopper circuit includes the boost inductor L101, the switching element Q101 which is for example an MOSFET, the rectifier diode D101, and the smoothing capacitor C104. The boost chopper circuit causes the switching element Q101 to go into the ON state or OFF state in response to the control signal F supplied from the AND circuit 127 of a control circuit section. Thereby, a boosted output voltage Vo is applied to both terminals of the smoothing capacitor C104.
The control circuit section includes a voltage sense operational amplifier 113, a multiplier 115, a current sense operational amplifier 117, the oscillator (OSC) 119, a PWM comparator 123, an inverter (INV) 121, an RS flip-flop circuit 125, and the AND circuit 127. The control circuit section is configured so as to supply the control signal F from the AND circuit 127 to the switching element Q101.
The boost chopper circuit and the control circuit section constitute a boost-chopper type active filter circuit. The active filter circuit carries out a PWM control such that a switching frequency of the switching element Q101 is fixed. Under such circumstances, in a case where input and output satisfy a predetermined condition, a direct current is superimposed on an electric current passing through the boost inductor L101. Such an active filter circuit is called a current continuous type active filter circuit.
The active filter circuit detects a waveform of an input voltage so as to cause an input current to be a sine wave similar to that of the input voltage. The waveform thus detected serves as a target value of a waveform of the input current which is the sine wave.
According to the example of FIG. 5, an input voltage from the smoothing capacitor C101 is divided by the resistor R101 and the resistor R102 which are connected in series with each other. The multiplier 115 receives a divided voltage via its input terminal C.
The output voltage Vo is divided by the resistor R108 and the resistor R109. The voltage sense operational amplifier 113 amplifies an error voltage between one of divided output voltages Vo and a reference voltage Vref. An amplified error voltage is supplied to an input terminal D of the multiplier 115 via a phase compensation circuit including the capacitor C106, the capacitor C107, and the resistor R107.
The multiplier 115 of FIG. 5 is a current output type multiplier. The multiplier 115 multiplies the amplified error voltage, which is supplied from the voltage detection operation amplifier 113, by a voltage supplied from a connection point between the resistor R101 and the resistor R102. The multiplier 115 then supplies an output signal E to an inverting input terminal (−) of the current sense operational amplifier 117. That is, the multiplier 115 determines a magnitude of a target sine wave current in accordance with a level of an error signal of the output voltage Vo (that is, in accordance with to what degree the output voltage Vo is different from the rectified voltage supplied from the bridge rectifier circuit DB).
The current sense operational amplifier 117 (i) compares the output signal E (a target value of a switching current) with the switching current detected by the resistor R104 (current sensing resistor) and (ii) amplifies an obtained result so as to obtain an output signal J, and then (iii) supplies the output signal J to an inverting input terminal (−) of the PWM comparator 123.
A capacitor CT is connected between the oscillator 119 and ground (GND). Similarly, the resistor RT is connected between the oscillator 119 and the ground (GND). The oscillation frequency of the oscillator 119 depends on a capacitance of the capacitor CT and a resistance of the resistor RT. The switching frequency of the switching element Q101 depends on the oscillation frequency of the oscillator 119.
The oscillator 119 repeats charging the capacitor CT and discharging electric charge stored in the capacitor CT, thereby generating a triangular wave signal A as illustrated in a timing chart of FIG. 6. Meanwhile, the oscillator 119 generates a square wave signal B on the basis of maximum and minimum values of the triangular wave signal A.
The triangular wave signal A is supplied to a non-inverting input terminal (+) of the PWM comparator 123. The square wave signal B is supplied to a reset terminal R of the RS flip-flop circuit 125 and an input terminal of the inverter 121.
The PWM comparator 123 supplies an H-level signal to a set terminal S of the RS flip-flop circuit 125 in a case where the triangular wave signal A from the oscillator 119 is equal to or larger than the output signal J from the current sense operational amplifier 117. On the other hand, the PWM comparator 123 supplies an L-level signal to the set terminal S of the RS flip-flop circuit 125 in a case where the triangular wave signal A from the oscillator 119 is smaller than the output signal J from the current sense operational amplifier 117.
The RS flip-flop circuit 125 outputs an H-level signal via its output terminal Q in response to the H-level signal received from the PWM comparator 123 via the set terminal S. The H-level signal thus outputted is supplied to a first input terminal of the AND circuit 127. On the other hand, the RS flip-flop circuit 125 is reset in response to the square wave signal B received from the oscillator 119 via the reset terminal R, and outputs an L-level signal via its output terminal Q. The L-level signal thus outputted is supplied to the first input terminal of the AND circuit 127.
The inverter 121 inverts the square wave signal B. An inverted square wave signal B (CLK with an overbar) is supplied to a second input terminal of the AND circuit 127.
The AND circuit 127 calculates AND of (i) the H-level or L-level signal supplied from the RS flip-flop circuit 125 via the output terminal Q and (ii) the inverted square wave signal B supplied from the inverter 121 so as to obtain a signal. The signal thus obtained, which signal serves as the control signal F, is supplied to the gate of the switching element Q101.
FIG. 6 illustrates a waveform of the control signal F. According to FIG. 6, the control signal F is in the H level during a period from a time t11 through a time t12 and a period from a time t14 through a time t15. Each of the time t11 and the time t14 is a time from which a level of the triangular wave signal A starts becoming higher than that of the output signal J. Each of the time t12 and the time t15 is a time from which a level of the square wave signal B starts changing from the L level to H level. This indicates that the square wave signal B in the H level is indicative of a start of a dead time during which the switching element Q101 is definitely in an OFF state.
According to the example of FIG. 6, a level of the output signal J from the current sense operational amplifier 117 increases as time passes. Along with this, a period during which the level of the control signal F, which is supplied to the gate of the switching element Q101, is being in the H level becomes shorter as time passes. Accordingly, the output voltage Vo becomes substantially constant as time passes. At the same time, a waveform of an electric current supplied to the switching power supply device 101 is controlled to be substantially a sine wave. This improves a power factor.
The fundamental wave component extraction circuit 118 includes the resistor R110 and the resistor R111, which are connected in series with each other between (i) the output terminal (i.e., the gate of the switching element Q101) of the AND circuit 127 and (ii) a connection point of the oscillator 119 and the resistor RT. The fundamental wave component extraction circuit 118 further includes the capacitor C105, which has the first terminal connected with the connection point of the resistor R110 and the resistor R111 and the second terminal electrically grounded.
The fundamental wave component extraction circuit 118 extracts the fundamental wave component of the control signal F for controlling the switching element Q101 through a CR filter constituted by the resistor R111 and the capacitor C105. The fundamental wave component extraction circuit 118 then supplies the control signal G, which is the fundamental wave component of the control signal F, to the resistor RT of the oscillator 119. Such extraction of the fundamental wave component of the control signal F is referred to as averaging.
FIG. 7, showing how the fundamental wave component extraction circuit 118 operates, is a timing chart illustrating signals from various sections. The following description discusses, with reference to FIG. 7, how the fundamental wave component extraction circuit 118 operates.
First, a rectified voltage is divided in the switching power supply device 101 of FIG. 5 for the purpose of causing an electric current, which is to be supplied to the switching power supply device 101, to be a sine wave. Specifically, the AC input voltage Vin supplied to the rectifier circuit DB via the filter 111 is rectified so as to be the rectified voltage, and then the rectified voltage is divided by the resistor R101 and the resistor R102. One of thus divided voltages is supplied to the input terminal C of the multiplier 115. Upon reception of the voltage, the control circuit section generates the control signal F for controlling the switching element Q101. That is, the control signal F contains a frequency component (i.e., sine wave component) of a commercial frequency of the AC input voltage Vin.
The fundamental wave component extraction circuit 118 extracts the fundamental wave component of the control signal F, thereby obtaining a voltage C105v of the capacitor C105 (see FIG. 7).
The voltage C105v of the capacitor C105 is applied to the resistor RT of the oscillator 119 via the resistor R110, which is a resistor for level adjustment. Thereby, the voltage C105v applied to the resistor RT changes as shown in FIG. 7. This makes it possible to cause the oscillation frequency of the oscillator 119 to change according to a change in the sine wave component of the commercial frequency. That is, it is possible to modulate a frequency in accordance with the sine wave component of the commercial frequency.
As a result, the switching frequency of the switching element Q101 changes within a certain range, and thus frequency components of a signal (PWM signal) supplied from the switching element Q101 are dispersed within a certain range. Accordingly, frequency components of noise, which is returned to an input terminal of the switching element Q101, are dispersed within a certain range according to the switching frequency. As a result, noise (voltage noise) having a frequency exists independently of other noise having another frequency. This prevents superimposition of noise, thereby reducing a voltage level of noise.
Note here that the noise returned to the input terminal of the switching element Q101 is noise caused by the recovery current.
Patent Literature 2 discloses a technique of reducing noise caused by switching like the switching power supply device 101 of FIG. 5, i.e., a DC-DC converter which reduces, at low cost, noise caused by a high-frequency component contained in a switching control signal.